Dynamic digital communication system control

ABSTRACT

Methods, apparatus and systems for dynamically controlling a digital communication system, such as a DSL system, collect information about digital communication lines in the system and adaptively and/or dynamically determine line and signal characteristics of the digital communication lines, including interference effects. Based on the determined characteristics and the desired performance parameters, operation of the digital communication lines is adjusted to improve or otherwise control the performance of the system. The collection and processing of information may be performed by a party that is not a user in the system. This independent party also may control operational characteristics and parameters of the system. The invention can be used to eliminated or reduce signal interference such as crosstalk that can be induced on communication lines in systems such as DSL systems. Specific iterative power allocation and vectored transmission techniques and apparatus are disclosed.

RELATED PATENT DOCUMENTS

This patent document is a continuation of U.S. patent application Ser.No. 09/877,724 filed on Jun. 8, 2001, and issued as U.S. Pat. No.7,158,563 to which priority is claimed under 35 U.S.C. §120; whichfurther claims benefit under 35 U.S.C. §119(e) of U.S. ProvisionalApplication Ser. No. 60/295,392 filed on Jun. 1, 2001, all of which arefull incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to methods, systems and apparatus formanaging digital communications systems. More specifically, theinvention relates to dynamically controlling system parameters thataffect performance in communication systems such as DSL systems.

2. Description of Related Art

The present invention refers to digital communication systems where thetransmission medium typically is copper wiring. Most commonly, thecopper wiring consists of twisted pairs (also referred to as “lines” or“loops”) categorized according to several manufacturing specifications(for example, AWG-26, AWG-24, CAT-3, CAT-5, CAT-6). Typicalcommunication systems making use of copper wiring include digitalSubscriber Line (DSL) systems, such as ISDN, HDSL, ADSL and VDSL, andLocal Area Networks (LAN), such as Ethernet. A transceiver (for example,a user modem) is situated at each end of the communications line thatincorporates the copper wiring.

Existing phone lines typically are “bundled” in some way. “Bundling”several pairs (in a binder or otherwise) can improve service to a singleuser or permit service for multiple users. For example, 1000-BaseTEthernet utilizes four twisted pairs to achieve a data rate of 250 Mbpsper pair, or an aggregate rate of 1 Gbps (shown in FIG. 1). In FIG. 1, adata stream 110 is fed to a first transceiver 120, where the data stream110 is decomposed into multiple component data streams 130 and, ifdesired modulated using a modulator 140. The modulated component datastream is transmitted over a twisted pair 150 to a demodulator 160 andre-composed in a second transceiver 170. Data may be sent in theopposite direction by reversing the roles of the various componentspreviously described.

Another application is the use of the telephone loop plant for DSLservice, one example of which is shown in FIG. 2. The twisted pairs 210emanating from each Customer Premises Equipment (CPE) 220 are groupedinto one or more binders 230, which converge at a terminus 240 such as acentral office (CO), an optical network unit (ONU), or a remote terminal(RT). Of course, hybrid scenarios may also occur, such as the use ofmultiple pairs by a single DSL customer aiming to improve his overalldata rate.

The bundling of twisted pairs arises either out of necessity (forexample, the existing telephone loop infrastructure) or because of thebenefits of improved performance (for example, 1000-BaseT Ethernet). Ineither case however, communications in these settings suffer frominterference arising from electromagnetic coupling between neighboringpairs, referred to as “crosstalk” interference. This means that anysignal received by a modem at the end of a twisted pair generallycontains not only the transmitted signal of the specific pair (whichitself is likely distorted to some extent), but also distorted signalstransmitted on neighboring pairs. It is apparent, therefore, that thetransmission characteristics of a specific pair (for example, the pair'stransmitted power) can materially influence communication on aneighboring pair due to the induced crosstalk. Therefore, transmissionson neighboring pairs (especially those belonging to a bundle or sharingthe same binder) are coupled in certain ways. The interfering signalsare commonly treated as noise. However, crosstalk can be identified insome situations. (See U.S. Ser. No. 09/788,267, now U.S. Pat. No.6,990,196 which is incorporated herein by reference.) If crosstalkcoupling functions can be identified, it may be possible to remove thecrosstalk interference.

“Unbundling” involves the incumbent local exchange carrier's (ILEC's)lease of a telephone line or some part of its bandwidth to a competitivelocal exchange carrier (CLEC). Current unbundling practice with DSLservice usually allows the CLEC to place modulated signals directly onleased physical copper-pair phone lines, sometimes referred to as thelease of “dark copper.” Such unbundled signals may provide services, andconsequently use spectra, that differ among the various serviceproviders. The difference in spectra can aggravate crosstalkingincompatibilities caused by electromagnetic leakage between linesexisting in close proximity. ILECs and CLECs try to ensure mutualspectral compatibility by standardizing the frequency bands and thepower spectral densities that can be used by various DSL services.However, there are many DSL types and bandwidths, and service providersare often competitors, which complicates such spectrum management.Further, the cooperation and connection between spectrum regulators andDSL standards groups is still in early evolution, so that regulators mayallow practices different than those presumed in spectrum management.

DSL spectrum management attempts to define the spectra of various DSLservices in order to limit the crosstalk between DSLs that may bedeployed in the same binder. Such crosstalk can be the limiting factorin determining the data rates and symmetries of offered DSL services atvarious loop reaches, so spectrum management finds some level ofcompromise between the various DSL service offerings that may besimultaneously deployed. Spectrum management studies tend to specifysome typical and worst-case loop situations, and then proceed to definefixed spectra for each type of DSL to reduce the mutual degradationbetween services. Such a fixed spectrum assignment may not produce thedesired level of compromise in situations different from those presumedin the studies.

These enacted rules place strict limits on transmission parameters,controlling performance degradation due to crosstalk by uniformlylimiting all parties' transmissions in the system. Typically, the entireset of rules applies equally irrespective of the actual crosstalkenvironment (for example, whether neighboring pairs actually transmitsignals or not), thereby providing protection for a worst-case scenario.

Currently, communication parameters at the physical layer (such astransmitted power, transmission bandwidth, transmitted power spectraldensity, energy allocation in time/frequency, bit allocation intime/frequency) are determined based on static information about a pairof modems and their twisted pair line. As seen in FIG. 3, an existingsystem 300 has modem pairs 310, 311 connected by twisted pair lines 312.Standardized requirements and constraints 314 for each link are providedto communication adaptation modules 315. In some cases measured line andsignal characteristics of a line 312 can be fed back to thecommunication adaptation module 315 by a module 316 for a given line toassist in operation of the modem pairs 310, 311 corresponding to theline 312. As illustrated in FIG. 3, however, there is no communicationor transfer of line and/or signal characteristics outside of each linkand its respective modem pair. Moreover, no independent entity hasknowledge of the operation of more than one modem pair and line or ofthe various pairs' interactions (for example, crosstalk between lines).Instead, the rules, requirements and constraints applied to lines andmodems such as those shown in FIG. 3 are designed to accommodate theworst cases of crosstalk or other interference, irrespective of theactual conditions present in the system during operation.

One of the shortcomings of current multi-user communication systems ispower control. In typical communication systems, which areinterference-limited, each user's performance depends not only on itsown power allocation, but also on the power allocation of all otherusers. Consequently, the system design generally involves importantperformance trade-offs among different users. The DSL environment can beconsidered a multi-user system, which would benefit from an advancedpower allocation scheme that maximizes or allows selection from most orall of the achievable data rates for multiple DSL modems in the presenceof mutual interference.

As mentioned above, DSL technology provides high speed data services viaordinary telephone copper pairs. The DSL environment is considered amulti-user environment because telephone lines from different users arebundled together on the way from the central office, and different linesin the bundle frequently create crosstalk into each other. Suchcrosstalk can be the dominant noise source in a loop. However, early DSLsystems such as ADSL and HDSL are designed as single-user systems.Although single-user systems are considerably easier to design, anactual multi-user system design can realize much higher data rates thanthose of single-user system designs.

As the demand for higher data rates increases and communication systemsmove toward higher frequency bands, where the crosstalk problem is morepronounced, spectral compatibility and power control are central issues.This is especially true for VDSL, where frequencies up to 20 MHz can beused.

Power control in DSL systems differs from power control in wirelesssystems because, although the DSL environment varies from loop to loop,it does not vary over time. Since fading and mobility are not issues,the assumption of perfect channel knowledge is reasonable. This allowsthe implementation of sophisticated centralized power control schemes.On the other hand, unlike the wireless situation where flat fading canoften be assumed, the DSL loops are severely frequency selective. Thus,any advanced power allocation scheme needs to consider not only thetotal amount of power allocated for each user, but also the allocationof power in each frequency. In particular, VDSL systems suffer from anear-far problem when two transmitters located at different distancesfrom the central offices both attempt to communicate with the centraloffice. When one transmitter is much closer to the central office thanthe other, the interference due to the closer transmitter oftenoverwhelms the signal from the farther transmitter.

DSL modems use frequencies above the traditional voice band to carryhigh-speed data. To combat intersymbol interference in the severelyfrequency selective telephone channel, DSL transmission uses DiscreteMultitone (DMT) modulation, which divides the frequency band into alarge number of sub-channels and lets each sub-channel carry a separatedata stream. The use of DMT modulation allows implementation ofarbitrary power allocation in each frequency tone, allowing spectralshaping.

As shown in FIG. 4, a DSL bundle 410 can consist of a number ofsubscriber lines 412 bundled together which, due to their closeproximity, generate crosstalk. Near-end crosstalk (NEXT) 414 refers tocrosstalk created by transmitters located on the same side as thereceiver. Far-end crosstalk (FEXT) 416 refers to crosstalk created bytransmitters located on the opposite side. NEXT typically is much largerthan FEXT. The examples of the present invention presented herein usefrequency duplexed systems for illustrative purposes.

Current DSL systems are designed as single-user systems. In addition toa system total power constraint, each user also is subject to a staticpower spectrum density (PSD) constraint. The power spectrum densityconstraint limits the worst-case interference level from any modem;thus, each modem can be designed to withstand the worst-case noise. Sucha design is conservative in the sense that realistic deploymentscenarios often have interference levels much lower than the worst-casenoise, and current systems are not designed to take advantage of thisfact. In addition, the same power spectrum density constraint is appliedto all modems uniformly regardless of their geographic location.

The absence of different power allocations for different users indifferent locations is problematic because of the near-far problemmentioned before. FIG. 5 illustrates a configuration in which two VDSLloops 510 in the same binder emanate from the central office 512 to afar customer premises 514 and a near customer premises 516. When bothtransmitters at the CPE-side transmit at the same power spectraldensity, the FEXT 526 caused by the short line can overwhelm the datasignal in the long line due to the difference in line attenuation. Theupstream performance of the long line is therefore severely affected bythe upstream transmission of the short line. To remedy this spectralcompatibility problem between short and long lines, the short lines mustreduce their upstream power spectral densities so that they do not causeunfair interference into the long lines. This reduction of upstreamtransmit power spectral density is known as upstream power back-off.Note that the downstream direction does not suffer from a similarproblem because, although all transmitters at the CO-side also transmitat the same power spectral density, the FEXT they cause to each other isidentical at any fixed distance from CO. This downstream FEXT level istypically much smaller than the data signals, so it does not pose aserious problem to downstream transmission.

Several upstream power back-off methods have been proposed in VDSL. Allcurrent power back-off methods attempt to reduce the interferenceemission caused by shorter loops by forcing the shorter loop to emulatethe behavior of a longer loop. For example, in the constant powerback-off method, a constant factor is applied across the frequency inupstream transmission bands, so that at a particular reference frequencythe received PSD level from shorter loops is the same as the receivedPSD level from a longer reference loop.

A generalization of this method is called the reference length methodwhere variable levels of back-off are implemented across the frequencyso that the received PSD for a shorter loop is the same as some longerreference loop at all frequencies. However, imposing the same PSD limitfor shorter loops across the entire frequency band may be toorestrictive since high frequency bands usually have too much attenuationto be useful in long loops. Therefore, short loops should be able totransmit at high frequency bands without worrying about theirinterference.

This observation leads to the multiple reference length method, whichsets a different reference length at each upstream frequency band. Allof the methods mentioned above equalize the PSD level of a shorter loopto the PSD level of some longer reference loop. While these methods maybe easy to implement in some cases, better performance can be obtainedif the interference levels themselves are equalized instead. Examples ofsuch approaches are the equalized-FEXT method, which forces the FEXTemission by shorter loops to be equal to the FEXT from a longerreference loop, and the reference noise method which forces the FEXTemission to equal to a more general reference noise. Although therepresently is no consensus on a single method, it is clear that aflexible method such as reference noise that allows spectrum shaping ismore likely to provide better performance.

Previously proposed power back-off methods require the power or noisespectrum of the short loops to comply with a reference loop or areference noise. These approaches are simple to implement because eachloop only needs to adjust its power spectrum according to a referenceand do not require any knowledge of the network configuration. If,however, loop and coupling characteristics in the network are known toeither the loops themselves, or a centralized third party, adaptiveadjustment power spectrum levels can be implemented, allowing bettersystem performance.

However, the optimization problem involved is complex as a result of thelarge number of variables and, due to the non-convex nature of theproblem, many local minima exist. Early attempts at solving this problemoften resorted to added constraints such as all transmitter powerspectrum densities being the same, or all PSDs being in some sensesymmetrical. The first attempt at finding the true global optimum isbased on quantum annealing to minimize the total energy subject to rateconstraints on each user.

As will be appreciated by those skilled in the art, the earlier methodsdescribed above had various shortcomings. Some of these methods weresimple to implement, but forfeited available performance for the sake ofsuch simplicity. The methods that attempted to realize higherperformance levels, on the other hand, were too complex to be practical.A relatively simple method and system that can achieve substantialimprovement in system performance would represent an importantadvancement in the art.

As noted above, DSL systems are rapidly gaining popularity as abroadband access technology capable of reliably delivering high datarates over telephone subscriber lines. The successful deployment ofAsymmetric DSL (ADSL) systems has helped reveal the potential of thistechnology. Current efforts focus on VDSL, which allows the use ofbandwidth up to 20 MHz. ADSL can reach downstream rates up to 6 Mbps,while VDSL aims to deliver asymmetric service with downstream rates upto 52 Mbps, and symmetric service with rates up to 13 Mbps. However, DSLcommunication is still far from reaching its full potential, and thegradual “shortening” of loops presents an opportunity to developadvanced methods that can achieve improved rates and performance.

In advanced DSL service the location of the line termination (LT or“central-office side”), as well as network termination (NT or “customerpremises side”), can vary. That is, not all LT modems are in the samephysical location. Often the location may be an ONU or cabinet, whereplacement and attachment of CLEC equipment may be technically difficultif not physically impossible. The difficulty arises because CLEC fiberaccess to the ONU may be restricted and/or the ONU may not be largeenough to accommodate a shelf/rack for each new CLEC. Placement of suchCLEC equipment for dark copper is often called “collocation” when it isin the central office. While space and facilitation of such centraloffice collocation for unbundling of the dark copper might be mandatedby law in some cases, an ILEC may only provide what is essentiallypacket unbundling at the LT terminal (that is, service bandwidth leasedat a layer 2 or 3 protocol level, not at the physical layer). Thisrepresents a change in the architecture presumed in many spectrumstudies.

Control of all the physical layer signals by a single service providerallows coordination of the transmitted signals in ways that can bebeneficial to performance of DSL service. Packet unbundling, which makesavailable the digital bandwidth on the twisted pairs, rather than thedirect physical layer lease of the line itself, is seen to be a likelystep in the evolution of DSL service.

A developing DSL system topology is shown in FIG. 6. Some twisted pairs616 emanate from the CO 610 and reach out to the customer premises 614.The installation of an ONU 612 (at a point between the CO 610 and one ormore CPEs 614) shortens loop lengths 618 so that the reach andperformance of DSL service are improved. Typically, the ONU 612 isconnected to the CO 610 through a fiber link 622. Pairs 616 and 618 canoccupy the same binder 620.

Crosstalk coupling is strongest among the twisted-pairs in a bindergroup. Therefore, eliminating or mitigating self-FEXT within a bindergroup has the biggest performance benefit. “Unbundled” lines ofdifferent service providers may share a binder group which can result inthe absence of collocation of the CO transceiver equipment. However,there are indications that ONU deployment will lead to an architecturein which some type of vectored transmission will be necessary sincedifferent service providers may have to “share” a fiber link to an ONU(for example, link 622 of FIG. 6) from which individual user lines willemanate and to which they will converge. More specifically, the currentarchitecture of “line unbundling” becomes impractical with theinstallation of ONUs, since line unbundling implies that each serviceprovider uses its own individual fiber to provide a proprietaryconnection to the ONU, and that the ONU must be large enough toaccommodate a shelf or rack for each service provider. Often, this isnot practical or possible. These difficulties may lead to the evolutionof “packet unbundling” where service bandwidth is leased at thetransport layer, instead of the physical layer. In that case, vectoredtransmission becomes more appealing because it can offer substantialperformance improvement and enhanced control.

The crosstalk problem has been addressed before with some shortcomings.For example, in some systems, MIMO Minimum-Mean-Square-Error (MMSE)linear equalizers were derived. Another prior method employs thesingular value decomposition to achieve crosstalk cancellation assumingco-location of both transmitters and receivers. Other earlier methodsinclude “wider than Nyquist” transmitters which were shown to provideperformance advantages compared to “Nyquist-limited” ones, and crosstalkcyclostationarity (induced by transmitter synchronization) combined withoversampling which were shown to result in higher SNR values.

None of the earlier methods or systems provided a relatively simple andeffective reduction in crosstalk interference in wireline communicationsystems. However, vectored transmission (as defined in this invention)can achieve a high degree of crosstalk reduction without unreasonablecomplexity. Moreover, the use of vectored transmission can accommodatethe approaching architectural changes coming to DSL service as well asproviding an opportunity for dynamic system management which canovercome the shortcomings of prior systems and methods.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to methods, apparatus and systems fordynamically controlling a digital communication system, such as a DSLsystem. Some embodiments of the present invention make use of some levelof knowledge regarding neighboring communication lines and/or thetransmission environment (for example, determining line and signalcharacteristics in one or more of a group of communication lines in thesystem) in order to improve performance on some or all of the lines.Generally, the present invention includes methods which determinephysical layer communication parameters, based on information obtainedabout the transmission environment, and which then evaluate optimizationcriteria related to corresponding links. Communication parameteradaptation may either occur once (for example, during modeminitialization), periodically or even continuously. The purpose ofperforming this joint adaptation is to utilize information about thechannel characteristics and about link requirements and constraints,which can yield improvements in the provisioning of services. Someembodiments of the present invention include methods where informationis gathered for all links, but the joint adaptation applies only to asubset of those links (even a single link).

Information regarding the line characteristics and signalingcharacteristics of all links can be shared or provided to an independententity in some embodiments. Line characteristics can include featuressuch as loop topology, transfer functions, and crosstalk couplingfunctions. For example, knowledge of crosstalk coupling can allowperformance improvements, since the amount of degradation of signals dueto transmission on one or more neighboring links can be accuratelyestimated. As a result, a change in system operation (for example, anincrease in transmitted power) might be determined to improveperformance without degrading neighboring links.

Signaling characteristics can include transmitted power spectraldensity, bandwidth utilized, modulation type, and bit allocation. Use ofthis information in connection with the present invention may allow thedistribution in frequency of the available power, so that the impactamong neighboring links is minimized.

In addition to the shared information regarding line and signalingcharacteristics, joint signal processing methods can be employed,utilizing knowledge of the transmitted bit streams. This coordinationlevel is directly related to the concept of “vectored” transmission,where crosstalk essentially is removed. Again, this allows a differentclass of adaptation methods, where the power and frequency resources ofall links can be optimally allocated in order to achieve the desiredrequirements.

More specifically, in one embodiment of the present invention, a digitalcommunication system is controlled by collecting information aboutdigital communication lines in the system and adaptively and/ordynamically determining line and signal characteristics of the digitalcommunication lines. Based on the determined characteristics and thedesired performance parameters, operation of the plurality of digitalcommunication lines is adjusted to improve or otherwise control theperformance of the digital communication system. The collection andprocessing of information may be performed by a party that is not a userin the system. This independent party also may control operationalcharacteristics and parameters of the system. In some embodiments, thepresent invention is used to eliminate or reduce signal interferencesuch as crosstalk that can be induced on groups of communication linesin systems such as DSL systems.

In another embodiment, a digital communication system has a number ofcommunication lines, each of the lines being used by a user, where thetotal power a user can use in the system is limited by a powerconstraint. A method of controlling system operation includes assigningthe total power constraint for each user an initial value and thendetermining a competitively optimal data rate for each user. Thecompetitively optimal data rate is ascertained by determining a powerallocation within the user's total power constraint as a result ofiteratively allowing each user to optimize its power allocation. Thecompetitively optimal data rate for a user is based on the powerallocation arrived at by that user and is evaluated by comparing thecompetitively optimal data rate with a target rate for the user.Depending on the comparison of the competitively optimal data rate andthe user's target rate, changes to the user's power constraint may bemade. The power constraint is increased for a user if the competitivelyoptimal data rate of the user is less than the target rate for the user;the power constraint is decreased if the competitively optimal data rateof the user exceeds the target rate for the user by at least aprescribed variance; and the power constraint remains the same if thecompetitively optimal data rate of the user is equal to the target ratefor the user or if the competitively optimal data rate of the userexceeds the target rate for the user by less than the prescribedvariance. This embodiment can be used in a line unbundling environment.

In another embodiment of the present invention, a digital communicationsystem has a plurality of communication lines on which signals aretransmitted and received, the signals being affected by interferenceduring transmission. Each of the communication lines is used by a userand has at least one transmitter and at least one receiver. A method ofcontrolling the system includes collecting information about line,signal and interference characteristics of the communication lines andcreating a mode of those line, signal and interference characteristicsof the communication lines. Transmission of signals between transmittersand receivers are synchronized to permit signal processing using themodel to remove interference from signals.

This embodiment can be used in a packet unbundling environment.

In another embodiment of the present invention, a method is implementedwhere communication signals are affected by interference duringtransmission on communication lines and each of the communication lineshas at least one transmitter and at least one receiver. A model iscreated of the interference characteristics due to the signals carriedon the communication lines.

Further details and advantages of the invention are provided in thefollowing Detailed Description and the associated Figures.

BRIEF DESCRIPTIONS OF THE DRAWINGS

The present invention will be readily understood by the followingdetailed description in conjunction with the accompanying drawings,wherein like reference numerals designate like structural elements, andin which:

FIG. 1 is a schematic diagram of a set of twisted pair telephone linesused for transmission of an aggregate information stream.

FIG. 2 is a schematic diagram of a DSL system utilizing an existingtelephone loop plant.

FIG. 3 is a schematic view of a communication system using linkrequirements and constraints and line and signal characteristicinformation on a per line basis.

FIG. 4 is a schematic representation of a DSL system showing a bundle oftransmission lines in a binder.

FIG. 5 is a schematic representation of the near-far problem encounteredwith FEXT crosstalk.

FIG. 6 is a schematic representation of a DSL system showing a bundle oftransmission lines in a binder wherein some of the lines share a fiberor other link between a CO and an ONU.

FIG. 7 is a schematic representation of one embodiment of the presentinvention in which information about line and signal characteristicsfrom a number of DSL lines is shared and used in a joint communicationadaptation configuration.

FIG. 8 is an interference channel model showing crosstalk interferenceamong DSL lines.

FIG. 9 is a timing diagram showing synchronization of block transmissionand reception at a CO/ONU.

FIG. 10 shows FEXT coupling measurements for loops with length of 1640feet.

FIG. 11 shows a canceller block of one embodiment of the presentinvention corresponding to a single tone in a discrete multitone system.

FIG. 12 shows a system for upstream vectored DMT transmission combiningcanceller blocks of all tones.

FIG. 13 shows a MIMO precoder of the present invention corresponding toa single tone in a discreet multitone system.

FIG. 14 shows a vectored DMT system for downstream transmissioncombining precoders of the present invention for all tones and includingthe DMT transmitter and receivers.

FIG. 15 illustrates the QR decomposition of two possible orderings.

FIG. 16 is a graphical depiction of differences in data rates inavailable with one embodiment of the present invention as a function ofloop lengths.

FIG. 17 is a flow diagram representation of one embodiment of thepresent invention in which power levels are determined.

DETAILED DESCRIPTION OF THE INVENTION

The following detailed description of the invention will be withreference to one or more embodiments of the invention, but is notlimited to such embodiments. The detailed description is intended onlyto be illustrative. Those skilled in the art will readily appreciatethat the detailed description given herein with respect to the Figuresis provided for explanatory purposes as the invention extends beyondthese limited embodiments. For example, the present invention isdescribed in some instances in connection with a DSL system. However,the present invention can be used with other systems that would benefitfrom the improved performance afforded by the present invention.Consequently, the present invention is not limited solely to DSLsystems. Moreover, the present invention is described herein primarilyin connection with the reduction of crosstalk interference. Again,however, the present invention may be used to reduce or eliminate otherundesirable signal interference or to otherwise improve the performanceof the system in which the present invention is used.

Performance of the system may be measured by maximizing data rates tousers. However, in some systems, operators may wish to be able to offera variety of services to users. For example, if an operator knows all ofthe available rates for a bundle, that operator may be able to offercertain users higher data rates as a “premium” service or forspecialized needs (such as a hospital or emergency care provider). Aswill be appreciated from the foregoing, terms such as “optimal” and“optimization” therefore may be subjectively defined and may notnecessarily refer to the fastest data rate(s), per se.

“Static spectrum management” uses fixed, inflexible constraints, limitsand requirements in connection with various digital communicationssystems. By contrast, a system with adaptive determination of spectra isreferred to herein as “dynamic spectrum management.” Necessarily, staticspectrum management is a special case of dynamic spectrum management, sostatic spectrum management can never outperform dynamic spectrummanagement. In fact, substantial improvement can be provided by dynamicspectrum management. The present invention illustrates that the level ofimprovement varies with loop characteristics, crosstalk couplingfunctions, data rates and symmetries offered, but can be significant.The level of relative improvement increases as loop lengths get shorterand data rates get more symmetric, as is likely to be the case with thepresent evolution of DSL. Importantly, dynamic spectrum managementaccording to the present invention allows a greater mix ofhigh-performance asymmetric and symmetric services in the same binder.

The present invention will be described in general with respect to adigital communications system. Within the context of dynamic spectrummanagement, however, there are two situations relating to the unbundlingof communications services that will be addressed by example inparticular—line unbundling and packet unbundling for DSL service. “Lineunbundling” occurs when different service providers place electricphysical-layer signals on copper wire lines within a telephone cable,which is the current practice when lines terminate in a central office.A specific illustrative example of the present invention (spectrumbalancing) will be presented below and is applicable in a lineunbundling environment. “Packet unbundling” occurs when serviceproviders instead lease bit streams from a single common carrier whomanages all signals on a telephone cable, meaning that different serviceproviders are utilizing the same telephone cable. This can occur, forexample, when fiber is used to connect a central office to an ONU, fromwhich different service providers' twisted pairs in turn emanate. Anillustrative example of the present invention (vectored transmission)will be explained below and is applicable in a packet unbundlingenvironment.

General

In some embodiments, the present invention uses methods which make useof some level of knowledge regarding the neighboring systems and thetransmission environment, in order to improve performance on all pairs.As a simple example, when crosstalk coupling between lines is weak,various transmission restrictions can be relaxed without substantialimpact. Going further, systems on neighboring pairs may shape theirpower spectral densities so that the mutually induced crosstalk isminimized and their performance targets are met.

The present invention further is defined to include methods andapparatus which determine and control physical layer communicationparameters, based on information obtained about the whole transmissionenvironment (the set of all neighboring twisted pairs) and whereoptimization criteria may relate to all the corresponding links. Thecommunication parameters also may refer to the time periods over whichtransmission on a pair is allowed, implying schemes similar to timedivision multiple access. The communication parameter adaptation canoccur once (for example, during modem initialization), periodically, oreven continuously.

The joint adaptation utilizes information about channel characteristicsand about link requirements and constraints, which results in theprovisioning of improved services. In some embodiments, information isgathered for all links, but the joint adaptation applies only to asingle subset of those links. In another embodiment of the presentinvention, information is gathered about all of the links, but the jointadaptation is applied independently to subsets of those links. In stillother embodiments, information may be gathered about only a subset ofthe links, with the joint adaptation being applied to all or a subset ofthe links.

One embodiment of the present invention is shown in FIG. 7. As withearlier systems, a digital communications system 700 uses pairs ofmodems 710, 711 which are connected by twisted pair lines 712. Universalrequirements and constraints (for example, total system power and powerconstraints on each line) can be applied to all links in the system by amodule 714. Again, line and signal characteristics for each line 712 canbe acquired and provided to the communication adaptation module 715. Theoperator of the module 715 may be a single service provider, a group ofservice providers or an independent entity 716 that collects andevaluates system data and provides instructions to users or, in somecases, possibly controls system parameters to achieve desirableoperational characteristics. In FIG. 7 the line and signalcharacteristics can be acquired for all (or a subset of) lines and canbe coordinated or otherwise considered in a joint manner.

In some embodiments of the present invention, information is sharedregarding the line characteristics of all links. One example can befound in U.S. Ser. No. 09/788,267, now U.S. Pat. No. 6,990,196 which isincorporated herein by reference. Line characteristics can include, butare not limited to, loop topology, transfer functions and crosstalkcoupling functions. For example, knowledge of crosstalk coupling canallow performance improvements, since the amount of degradation of alink due to transmission on a neighboring link can be accuratelyestimated, and thus it may be realized that an increase in thetransmitted power will improve the performance of the link withoutdegrading the neighboring links.

In still other embodiments of the present invention, information also(or instead) is shared regarding the signaling characteristics.Signaling characteristics can include, but are not limited to,transmitted power spectral density, bandwidth utilization andallocation, modulation type and bit allocation. This may allow theapplication of a new class of methods and apparatus, such as thoseinvolving the distribution in frequency of available power, so that theimpact among neighboring links is minimized.

In addition to sharing information regarding line and/or signalingcharacteristics, joint signal processing methods can be employed whichwill utilize knowledge of the transmitted bit streams. This coordinationlevel is directly related to the concept of “vectored” transmission,where crosstalk is essentially removed. Again, this allows a differentclass of adaptation methods, where the power and frequency resources ofall links can be optimally allocated in order to achieve the desiredrequirements.

Two specific implementations of the present invention are now presented.The first uses an adaptive multi-user power control methodology,presented as an example as applied to a VDSL system. Such a system isuseful in a line unbundling environment where different serviceproviders may have access to different lines in a binder and/ordifferent services that potentially negatively affect one another areprovided on the lines in the binder.

Adaptive Power Control Method

The digital subscriber line (DSL) environment can be viewed as amulti-user system. One embodiment of the present invention is intendedto optimize power allocation to identify the maximum achievable datarates for multiple DSL modems in the presence of mutual interference.The following discussion will use VDSL as an example, and show that amulti-user system design with an advanced power allocation scheme canprovide a system with substantial performance improvement as compared toa single-user design that does not take the multi-user aspect intoaccount. This advanced power allocation method can be implemented eitherin a centralized fashion or a distributed fashion. The centralizedapproach assumes the existence of an entity which acquires knowledge ofchannel and crosstalk coupling functions, determines the desiredsignaling characteristics and parameters for each user, and finallyinstructs each user to employ these transmission characteristics andparameters.

Another embodiment does not require knowledge of the crosstalk couplingfunctions. In such an embodiment, the modems of each user enter a phaseduring which each user individually adjusts its own signalingcharacteristics with the aim of attaining its own desired performancelevel, while minimizing the crosstalk it induces on the other users. Inthis embodiment, a centralized entity may still exist, but its role maybe restricted to setting the target performance levels of each user.

The following discussion will evaluate transmission techniques where nomulti-user detection takes place, and focus solely on advanced powerallocation for each user in the network. An interference channel model800 is shown in FIG. 8. There are N transmitters 810-1 through 810-N andN receivers 820-1 through 820-N in the network 800. The channel fromuser i to user j is modeled as an ISI channel, whose transfer functionin frequency domain is denoted as H_(ij)(ƒ), where

${0 \leq f \leq F_{s}},{F_{s} = \frac{1}{2\; T_{s}}},$and T_(s) is the sampling rate. In addition to the interference noise,each receiver also sees a background noise whose power spectrum densityis denoted as σ_(i)(ƒ). The power allocation for each transmitter isdenoted as P_(i)(ƒ), which must satisfy a power constraint:∫₀ ^(F) ^(s) P _(i)(ƒ)dƒ≦P _(i)  Equation (1)The achievable data rate for each user while treating all interferenceas noise is:

$\begin{matrix}{R_{i} = {\int_{0}^{F_{s}}{{\log_{2}\left( {1 + \frac{{P_{i}(f)}{{H_{ii}(f)}}^{2}}{\Gamma\left( {{\sigma_{i}(f)} + {\sum\limits_{j \neq i}{{P_{j}(f)}{{H_{ji}(f)}}^{2}}}} \right)}} \right)}{\mathbb{d}f}}}} & {{Eq}.\mspace{14mu}(2)}\end{matrix}$where Γ denotes the SNR-gap which depends on the probability of error,the modulation scheme and the coding applied. A coding and modulationscheme that approaches the information theoretical capacity has Γ=0 dB.

The objective of the system design is to maximize the set of rates {R₁,. . . , R_(N)} subject to the power constraints of Equation (1). It willbe apparent to those of skill in the art that, for each transmitter,increasing its power at any frequency band will increase its own datarate. However, such an increase also causes more interference to otherusers and is therefore detrimental to other users' transmissions. Thus,an optimization or other advanced design must consider the trade-offamong the data rates of all users.

Realistic DSL deployment often requires multiple service rates besupported for all users, and the required level of service of each usercould be arbitrary. Therefore, a single figure of merit frequently isinadequate to represent system performance. Also, as noted above, onemay wish to know all achievable data rate combinations for the users ina system. For example, if the objective is to maximize the sum rate,then there is no guarantee of a minimal data rate to any one user.

A convenient way to fully characterize the trade-off among the users andthe achievable data rates available to them is through the notion of arate region, which is defined as:R={(R ₁ , . . . , R _(N)):∃(P ₁(ƒ), . . . , P _(N)(ƒ)) satisfying Eqs.(1) and (2)}.  Eq. (3)

The rate region characterizes all possible data rate combinations amongall users. Although in theory, the rate region can be found by anexhaustive search through all possible power allocations, or by a seriesof optimizations involving weighted sums of data rates, computationalcomplexities of these approaches typically are prohibitively high. Thisis because the rate formula is a non-convex function of powerallocations. Consequently, the usual numerical algorithms are able tofind only local maxima and not the global maximum. The present inventionavoids these complexities by defining a different concept of competitiveoptimality. Although the methodology of this embodiment of the presentinvention does not achieve all points in the rate region defined above,it nevertheless performs much better than the current DSL systems.

Instead of finding the global maximum, the present invention utilizescompetitive optimality, which has the advantage of providing the locallyoptimal solution toward which all users have an incentive to move. Thesecompetitively optimal points are easy to characterize, and they lead toa power control method that offers a number of advantages compared tothe previous methods. First, unlike previous methods that set a PSDlevel for each VDSL transmitter based solely on its interferenceemission level, the new power allocation method of the present inventionstrikes a balance between maximizing each user's own data rate andminimizing its interference emission. In particular, the frequencyselective nature of the channel is dealt with explicitly. Second, bytaking into account all loop transfer functions and cross-couplings(directly in the embodiment using a centralized control entity, tacitlyin the embodiment using a distributed method), the method of the presentinvention offers the loops an opportunity to negotiate the best use ofpower and frequency with each other. Third, the usual PSD constraint,which is in place for the purpose of controlling interference, is nolonger needed. Only total power constraints apply. Fourth, unlikeprevious methods, which fix a data rate for each loop regardless ofactual service requirement, the new method naturally supports multipleservice requirements in different loops. Fifth, the proposed method doesnot involve arbitrary decisions on the reference noise or referencelength. Finally, much better performance can be achieved both in termsof maximum data rates and selectivity of services and/or rates within asystem.

Competitive Optimality

The traditional information-theoretic view of an interference channelallows the different transmitters, while sending independent datastreams, to be cooperative in their respective coding strategies, sothat interference cancellation can take place in the receivers. If suchcooperation cannot be assumed, the interference channel can be bettermodeled as a non-cooperative game. Under this viewpoint, each usercompetes for data rates with the sole objective of maximizing itsperformance, regardless of all other users. This scenario isparticularly realistic in the current unbundled environment wheredifferent loops in the same binder could belong to different serviceproviders, and they indeed compete in the local access market. Now,because each modem has a fixed power budget, each user should adjust itspower allocation to maximize its own data rate, while regarding allother interference as noise.

If such power adjustment is done continuously for all users at the sametime, they will eventually reach an equilibrium. Such an equilibriumwill be a desired system operating point since, at equilibrium, eachuser will have reached its own local maximum, and nobody has anincentive to move away from that power allocation. From a game theoryperspective, this equilibrium point is called the Nash equilibrium.

A Nash equilibrium is defined as a strategy profile in which eachplayer's strategy is an optimal response to each other player'sstrategy. The following discussion will characterize the Nashequilibrium in the Gaussian interference channel game, and determine itsexistence and uniqueness in realistic channels.

A two-user interference channel provides the following simplified model:y ₁ =x ₁ +A ₂ x ₂ +n ₁  Eq. (4)y ₂ =x ₂ +A ₁ x ₁ +n ₂  Eq. (5)where the channel transfer functions are normalized to unity. The squaremagnitude of the interference transfer functions A₁ and A₂ are denotedas α₁(ƒ) and α₂(ƒ), respectively. Let N₁(ƒ) and N₂(ƒ) denote noise powerspectrum densities. The two senders are considered as two players in agame. The structure of the game (that is, the interference couplingfunctions and noise power) are assumed to be common knowledge to bothplayers. A strategy for each player is its transmit power spectrum,P₁(ƒ) and P₂(ƒ), subject to the power constraints ∫₀ ^(F) ^(s)P₁(ƒ)dƒ≦P₁ and ∫₀ ^(F) ^(s) P₂(ƒ)dƒ≦P₂, respectively, considering onlydeterministic, or pure strategy here. The payoff for each user is itsrespective data rate. Under the simplifying assumption that nointerference subtraction is performed regardless of interferencestrength, the data rates are:

$\begin{matrix}{R_{1} = {\int_{0}^{F_{s}}{{\log\left( {1 + \frac{P_{1}(f)}{{N_{1}(f)} + {{\alpha_{2}(f)}{P_{1}(f)}}}} \right)}{\mathbb{d}f}}}} & {{Eq}.\mspace{14mu}(6)} \\{R_{2} = {\int_{0}^{F_{s}}{{\log\left( {1 + \frac{P_{2}(f)}{{N_{2}(f)} + {{\alpha_{1}(f)}{P_{2}(f)}}}} \right)}{\mathbb{d}f}}}} & {{Eq}.\mspace{14mu}(7)}\end{matrix}$If we compare the above expression with equation (2), we can identify:

$\begin{matrix}{{{N_{1}(f)} = \frac{\Gamma\;{\sigma_{1}(f)}}{{{H_{11}(f)}}^{2}}}{and}} & {{Eq}.\mspace{14mu}(8)} \\{{{\alpha_{2}(f)} = \frac{{{H_{21}(f)}}^{2}}{\Gamma{{H_{11}(f)}}^{2}}}{and}} & {{Eq}.\mspace{14mu}(9)} \\{{{N_{2}(f)} = \frac{\Gamma\;{\sigma_{2}(f)}}{{{H_{22}(f)}}^{2}}}{and}} & {{Eq}.\mspace{14mu}(10)} \\{{\alpha_{1}(f)} = \frac{{{H_{12}(f)}}^{2}}{\Gamma{{H_{22}(f)}}^{2}}} & {{Eq}.\mspace{14mu}(11)}\end{matrix}$Thus, the simplified model incurs no loss of generality.

The data rate game discussed here is not a zero-sum game. That is, oneplayer's loss is not equal to the other player's gain. Since at a Nashequilibrium, each user's strategy is the optimal response to the otherplayer's strategy, and for each user, the optimal power allocation givenother player's power level is the water-filling of the power against thecombined noise and interference, a Nash equilibrium is reached ifwater-filling is simultaneously achieved for all users.

A complete characterization of the simultaneous water-filling point inthe interference channel may be difficult to do, however there areseveral sufficient conditions for the existence and uniqueness of theNash equilibrium in the two-user case. For all α₁(ƒ)α₂(ƒ)<1, ∀ƒ, then atleast one pure strategy Nash equilibrium in the Gaussian interferencegame exists. Further, if:

$\begin{matrix}{\lambda_{0} = {\sup\left\{ {\alpha_{1}(f)} \right\}\sup\left\{ {\alpha_{2}(f)} \right\}}} & {{Eq}.\mspace{14mu}(12)} \\{\lambda_{1} = {\sup\left\{ {{\alpha_{1}(f)}{\alpha_{2}(f)}} \right\}}} & {{Eq}.\mspace{14mu}(13)} \\{\lambda_{2} = {\sup\left\{ {\alpha_{1}(f)} \right\}\frac{1}{F_{s}}{\int_{0}^{F_{s}}{{\alpha_{2}(f)}{\mathbb{d}f}}}}} & {{Eq}.\mspace{14mu}(14)} \\{{\lambda_{3} = {\sup\left\{ {\alpha_{2}(f)} \right\}\frac{1}{F_{s}}{\int_{0}^{F_{s}}{{\alpha_{1}(f)}{\mathbb{d}f}}}}}{{and}\mspace{14mu}{either}}} & {{Eq}.\mspace{14mu}(15)} \\{{\lambda_{0} < 1},{or}} & {{Eq}.\mspace{14mu}(16)} \\{{{\lambda_{1} + \lambda_{2}} < \frac{1}{2}},\;{or}} & {{Eq}.\mspace{14mu}(17)} \\{{\lambda_{1} + \lambda_{3}} < \frac{1}{2}} & {{Eq}.\mspace{14mu}(18)}\end{matrix}$then the Nash equilibrium is unique and is stable.

The convergence of the iterative water-filling process shows that theNash equilibrium is unique. This is because the starting point isarbitrary, so it could be another Nash equilibrium if it were notunique. But each Nash equilibrium is its own fixed point. So, thiscannot happen. The stability of the Nash equilibrium also follows fromthe convergence of the iterative procedure.

Moreover, if the condition for existence and uniqueness of the Nashequilibrium is satisfied, then an iterative water-filling algorithm,where in every step each modem updates its power spectrum densityregarding all interference as noise, converges and it converges to theunique Nash equilibrium from any starting point.

Adaptive Power Control

Because of the frequency-selective nature of the DSL channel and the DSLcrosstalk coupling, power control algorithms in the DSL environment notonly need to allocate power among different users, they also need toallocate power in the frequency domain. This requirement brings in manyextra variables and makes the design of advanced power control for DSLdifficult. However, by concentrating on the competitive optimal powerallocations, and assuming that the existence and uniqueness conditionsunder a total power constraint for Nash equilibrium are satisfied, totalpower alone is sufficient to represent all competitive powerallocations. Consequently, power control can be based solely on totalpower despite the frequency selectivity. This simplifies the processtremendously. Although the competitive optimal solutions are in generalnot globally optimal, impressive improvement can still be realized whencompared to existing power back-off algorithms.

The goal is to achieve certain target rates for each user. The adaptiveprocess runs in two stages. The inner stage uses given power constraintsfor each user as the input and derives the competitive optimal powerallocation and data rates as output. This is accomplished by theiterative water-filling procedure. With a fixed total power constraintfor each user, the first user updates its power allocation as thewater-filling spectrum of its channel regarding all other users'crosstalk as noise. Water-filling is then applied successively to thesecond user, the third user, and so forth, then again to the first user,second user, etc., until each user's power allocation converges.Alternative (or even random) orderings also will work, provided that allusers are “served” in due course.

The outer stage finds the optimal total power constraint for each user.The outer procedure adjusts each user's power constraint based on theoutcome of the inner iterative water-filling. If a user's data rate isbelow the user's target rate, then the user's power constraint will beincreased, unless it is already at the modem power limit, in which caseits power stays the same. If a user's data rate is above its target rateby a prescribed amount, its power will be decreased. If the data rate isonly slightly above the target rate (less than the prescribed amount),its power will be unchanged. The outer procedure converges when the setof target rates is achieved.

The method described above applies to the distributed version, whereeach user acts independently, apart from the fact that its target datarate has been “imposed” on the user by an outside agent or entity. It iseasy to derive a centralized version, where a central entity performsthe inner and outer iteration steps, and then determines powerallocations, which it then instructs the users to adopt. The centralizedversion implies that the entity has acquired knowledge of some or all ofthe line and/or signal characteristics.

In a K-user system, using P as the modem power limit and T_(i) as thetarget rate of user i, the preferred process can be summarized asfollows:

-   Initialize P_(i)=P, i=1, . . . , K.-   repeat    -   repeat        -   for i=1 to K

${{N(f)} = {{\sum\limits_{{j = 1},{j \neq i}}^{K}{{{H_{ji}(f)}}^{2}{P_{j}(f)}}} + {\sigma_{i}(f)}}};$

-   -   -   -   P_(i)(ƒ)=water-filling spectrum with channel                |H_(ii)(ƒ)|², noise N(ƒ), and power constraint P_(i)            -   R_(i)=data rate on channel |H_(ii)(ƒ)|² with the power                allocation P_(i)(ƒ), and noise N(ƒ)

        -   end

    -   until the desired accuracy is reached.

    -   for i=1 to K        -   If R_(i)>T_(i)+ε, set P_(i)=P_(i)−δ.        -   If R_(i)<T_(i), set P_(i)=P_(i)+δ.        -   If P_(i)>P, set P_(i)=P.

    -   end

-   until R_(i)>T_(i) for all i.

This process works well with the parameters δ=3 dB and ε equal to 10% ofthe target rate. The outer iteration converges only if the set of targetrates is achievable. Unfortunately, which set of target rates isachievable cannot be known α priori. However, a centralized agent withfull knowledge of all channel and interference transfer functions candecide, by running through all possible total power constraints, whichsets of target rates can be deployed in a DSL bundle. In effect, thepower allocation problem has been separated into two parts. Acentralized agent may decide on a target rate and a power constraint foreach loop in the bundle. Then the loops themselves can undergo theiterative water-filling procedure to arrive at the desired rates withoutthe need of centralized control. The amount of information that needs tobe passed by the central control to each loop is small.

Comparing the present invention to conventional power control methods,this new method offers two key advantages. First, because theinterference is controlled systematically, no power spectral densityconstraints are needed, thus, allowing more efficient use of total powerby all users. Secondly, because a single-user water-filling is performedat each stage, optimizing each user's data rate regarding all otherusers as noise, the iterative water-filling algorithm offers anopportunity for different loops in a binder to negotiate the use offrequency. Thus, each loop has an incentive to move away from frequencybands when interference is strong, and concentrate on the frequencybands that it can most efficiently utilize.

Simulations show that the competitively optimal power allocation methodof the present invention offers a dramatic improvement in performance.This improvement is possible because the new power control methodologyconsiders all loops in the binder as a whole, taking into account allinteractions and globally allocating power to each user. Although thecompetitively optimal operating points are not necessarily globallyoptimal, the present invention offers substantial improvement overcurrent power back-off methods which consider only each loop by itself.These competitively optimal points are easy to find because iterativewater-filling converges very fast.

It should be noted that other “transmission optimization” techniques andmethods can be used instead of the disclosed water-filling method (forexample, discrete loading methods). Also, the rate maximizationcriterion can be replaced by a margin maximization criterion, where thetarget data rates are fixed for each user.

Vectored Transmission

In the following example, vectored transmission for DSL systems isexplained. This implementation of the present invention is useful in apacket unbundling environment where a single line is used by multipleusers (for example, when leased by a single operator or where a fiberconnection ends at an ONU and provides multiple parties with servicefrom multiple service providers).

Channel Model and DMT Transmission

The DSL channel model for the architecture of FIG. 4 is now presented.The L users 420-1 through 420-L are assumed to correspond to a subset ofthe twisted-pairs of a group in binder 410. The sampled output for aspecific user for either upstream or downstream transmission depends onthe present and past input symbols of both the intended user and theother crosstalking users. A block of N output samples for user isatisfies:y _(i) =H _(i,1) ^(c) x ₁ ^(p) + . . . +H _(i,i) ^(c) x _(i) ^(p) + . .. +H _(i,L) ^(c) x _(L) ^(p) +n _(i)  Eq. (19)where H_(i,1) ^(c), . . . , H_(i,L) ^(c) are convolution matricesderived from the channel impulse response matrix, y_(i) is the vector ofN output samples of receiver i, x_(k) ^(p) is the vector of N+ν inputsymbols of user k, and n_(i) is the vector of N noise samples ofreceiver i. ν represents the maximum memory of the transfer andcrosstalk coupling functions expressed in number of samples. The noisesamples represent the superposition of several noise sources such ascrosstalk from neighboring DSL systems, radio frequency ingress andimpulse noise. In the following, n_(i) is considered to be white andgaussian and, without loss of generality, has unit variance.

Two fundamental assumptions are used in connection with this discussionof a preferred embodiment. First, all users employ block transmissionwith a cyclic prefix (CP) of at least length ν. Also, block transmissionand reception at the CO/ONU are synchronized as illustrated in thetiming diagram of FIG. 9.

Given the co-location assumption for the CO/ONU, synchronized blocktransmission is relatively straightforward to implement. However,synchronized block reception requires additional consideration, althoughvarious methods and configurations will be apparent to those of ordinaryskill in the art. The block boundaries for upstream transmission arealigned so that the blocks of all users arrive simultaneously at theCO/ONU. This block-level synchronization can be performed duringinitialization, and is analogous to the problem of synchronized uplinktransmission in a wireless environment.

Synchronization at the CO/ONU is automatically achieved when “zipper”FDD is used. According to this technique, a cyclic suffix (CS) largerthan the channel propagation delay is included in addition to the CP.This “zippering” offers the benefit of eliminating residual NEXT andnear-echo resulting from “spectral leakage” at frequencies close to theupstream/downstream band edges. Nevertheless, in this disclosure, theless stringent assumptions noted above will be used, understanding thatresidual NEXT and near-echo are mitigated by transmitter pulse-shapingand receiver windowing which are known in the art.

Taking the above into account, Equation (19) becomes:y _(i) =H _(i,1) x ₁ + . . . +H _(i,i) x _(i) + . . . +H _(i,L) x _(L)+n _(i)  Eq. (20)where x_(k) is a vector of N input symbols of user k, and H_(i,j),i,j=1, . . . , L are circulant matrices. Combining the L users, Equation(20) becomes:y=Hx+n  Eq. (21)where y=[y₁ ^(T)y₂ ^(T) . . . y_(L) ^(T)]^(T), x=[x₁ ^(T)x₂ ^(T) . . .x_(L) ^(T)]^(T), n=[n₁ ^(T)n₂ ^(T) . . . n_(L) ^(T)]^(T), and H is amatrix whose (i,j) block is H_(i,j). The noise covariance matrix isassumed to be R_(nn)=I.

Applying Discrete Fourier Transform (DFT) modulation, which is known inthe art, an Inverse Discrete Fourier Transform (IDFT) operation isperformed on each transmitted data block (prior to appending the CP),and a DFT operation is performed on each received data block (afterdiscarding the CP), thus yielding a channel description where thesamples are stacked in groups corresponding to users, and each of thegroups contains samples corresponding to tones. It is desirable toreorganize these samples for further processing by stacking in groupscorresponding to tones, where each group contains samples correspondingto different users. To this end, a permutation matrix P having NL rowsand NL columns is defined, which is composed of the N×N blocks P_(i,j)where i,j=1, . . . , L. The block P_(i,j) contains all zeros, except fora one at position (j,i). When matrix P is right-multiplied with a vectorof size NL, the elements of P are re-ordered from L groups of Ncomponents into N groups of L components. Also, note that P⁻¹=P*=P.Applying this reordering operation to both the transmitter and thereceiver samples, yields:Z _(i) =T _(i) U _(i) +N _(i) , i=1, . . . , N  Eq. (22)

Therefore, Z_(i), U_(i) and N_(i) contain the received samples,transmitted symbols and noise samples of all users corresponding to tonei, and T_(i) fully characterizes MIMO transmission within tone i. In thefollowing, a distinction between upstream and downstream will be made byadopting the notation T_(i,up) and T_(i,down).

Equation (22) shows that crosstalk cancellation can be performedindependently in each tone. Therefore, as explained in more detailbelow, an array of canceller blocks can be employed at the CO/ONU toremove crosstalk within each tone for upstream communication. Similarly,precoder blocks can be used at the CO/ONU to pre-distort the transmittedsignals within each tone, so that signals received at the CPE arecrosstalk-free. Determining the parameters of the canceller/precoderblocks relies on perfect knowledge of the channel matrix and noisecovariance matrix at the CO/ONU. This assumption is reasonable for DSL,since the twisted pair channels are stationary, and systems can affordtraining-based channel identification during initialization.

The additional requirement of having a CP longer than the memory of boththe transfer and the crosstalk coupling functions can be satisfiedwithout suffering an excessive loss. FIG. 10 shows FEXT couplingmeasurements for loops with length of 1640 feet. Since only magnitudedata is provided, linear phase was assumed in order to derive theimpulse responses. It was found that 99.9% of the signal energy iscontained within 9 sec. With a DMT block size of 4096 samples andsampling rate of 17.664 MHz, this corresponds to 159 samples. Therefore,a CP length of 320 samples (corresponding to a 7.8% loss) is more thanadequate.

The average delay of a typical twisted pair is approximately 1.5 μs/kft.Given that VDSL loops usually have lengths shorter than 6000 ft, andwith the previous DMT assumptions, the propagation delay corresponds tofewer than 160 samples. Therefore, even if “zippering” is used, thelength of the CP plus the CS does not exceed the proposed 320 samples.As is known to those of ordinary skill in the art, in cases where thechannel has unusually long memory, various techniques are available for“shortening” the memory. For example, a MIMO Time-Domain-Equalizer maybe used at the CO/ONU and a MIMO extension of an appropriate precodermay be utilized for downstream communication.

Crosstalk Cancellation Via QR Decomposition

Starting with Equation (22), the methods to remove crosstalk within eachtone are described first for upstream and then for downstreamcommunication. In the following, the matrices T_(i,up) and T_(i,down)are assumed to be non-singular (the justification for this assumptionand the consequences of ill-conditioning are discussed below).

Upstream

For upstream transmission, the co-location of the CO/ONU transceiverequipment gives the opportunity to perform joint signal processing ofthe received samples. The computation of the QR decomposition of matrixT_(i,up) yields:T _(i,up) =Q _(i) R _(i)  Eq. (23)where Q_(i) is a unitary matrix and R_(i) is an upper triangular matrix.If the received samples are “rotated/reflected” by Q_(i)*, then Equation(22) becomes:

$\begin{matrix}{{\overset{\sim}{Z}}_{i} = {Q_{i}^{*}\; Z_{i}}} & {{Eq}.\mspace{14mu}(24)} \\{\mspace{25mu}{= {{R_{i}U_{i}} + {\overset{\sim}{N}}_{i}}}} & {{Eq}.\mspace{14mu}(25)}\end{matrix}$where Ñ_(i)=Q_(i)*N_(i) has an identity covariance matrix. Since R_(i)is upper triangular and Ñ_(i) has uncorrelated components, the inputU_(i) can be recovered by back-substitution combined withsymbol-by-symbol detection. Thus, as seen in FIG. 11, a decisionfeedback structure 1100 is created with the feedforward matrix module1110 using Q_(i)*, and the feedback matrix module 1120 using I−R_(i).The detection of the k^(th) element of U_(i) is expressed as:

$\begin{matrix}{{\left( \overset{\sim}{U} \right)_{k} = {{dec}\left\lbrack {{\frac{1}{r_{k,k}^{i}}\left( {\overset{\sim}{Z}}_{i} \right)_{k}} - {\sum\limits_{j = {k + 1}}^{L}{\frac{r_{k,j}^{i}}{r_{k,k}^{i}}\left( \hat{U} \right)_{j}}}} \right\rbrack}},{k = L},{L - 1},\ldots\mspace{14mu},1} & {{Eq}.\mspace{14mu}(26)}\end{matrix}$where r_(k,j) ^(i) is the (k,j) element of R_(i). Assuming that theprevious decisions are correct, crosstalk is completely cancelled, and L“parallel” channels are created within each tone. The operationsdescribed above can be used to define a preferred canceller blockcorresponding to a single tone, which is shown in FIG. 11. Combining thecanceller blocks of all tones, and taking into account DMT transmission,a system 1200 for upstream vectored DMT transmission is shown in FIG.12. The transmitters 1210-1 through 1210-L send their respective signalsthrough channel 1220. The receivers 1230-1 through 1230-L receive thesignals from channel 1220 and process the received signals usingcanceller blocks 1240-1 through 1240-L which, in the preferredembodiment, resemble the block of FIG. 11.Downstream

For downstream transmission in the preferred embodiment, joint signalprocessing of the transmitted symbols is used. The QR decomposition ofT_(i,down) ^(T) results in:T _(i,down) ^(T) =Q _(i) R _(i)  Eq. (27)where again Q_(i) is a unitary matrix and R_(i) is an upper triangularmatrix. Assuming that the symbols are “rotated/reflected” by Q_(i) ^(T)*prior to being transmitted:U _(i) =Q _(i) ^(T) *U′ _(i)  Eq. (28)So, choosing:

$\begin{matrix}{{\left( U_{i}^{\prime} \right)_{k} = {\Gamma_{M_{i,k}}\left\lbrack {\left( {\overset{\sim}{U}}_{i} \right)_{k} - {\sum\limits_{j = 1}^{k - 1}{\frac{r_{j,k}^{i}}{r_{k,k}^{i}}\left( U_{i}^{\prime} \right)_{j}}}} \right\rbrack}},{k = 1},2,\ldots\mspace{14mu},L} & {{Eq}.\mspace{14mu}(29)}\end{matrix}$crosstalk-free reception is achieved, where the transmitted symbols intone i are the elements of Ũ_(i). The following operation is performedat the receiver:

$\begin{matrix}{{\left( {\hat{Z}}_{i} \right)_{k} = {\Gamma_{M_{i,k}}\left\lbrack \frac{\left( Z_{i} \right)_{k}}{r_{k,k}^{i}} \right\rbrack}},{k = 1},2,\ldots\mspace{14mu},L} & {{Eq}.\mspace{14mu}(30)}\end{matrix}$where Γ_(M) _(i,k) is defined as:

$\begin{matrix}{{\Gamma_{M_{i,k}}\lbrack x\rbrack} = {x - {M_{i,k}{d\left\lbrack \frac{x + \frac{M_{i,k}d}{2}}{M_{i,k}d} \right\rbrack}}}} & {{Eq}.\mspace{14mu}(31)}\end{matrix}$and M_(i,k) is the constellation size of user k on tone i, while d isthe constellation point spacing.

$\left. {{\Gamma_{M_{l,k}}\lbrack x\rbrack} = {{\Gamma_{\sqrt{M_{l,k}}}\left\lbrack {(x)} \right\rbrack} + {j\;{{\Gamma_{\sqrt{M_{l,k}}}\left\lbrack {(x)} \right\rbrack}.}}}} \right)$These operations result in:{circumflex over (Z)} _(i) =Ũ _(i)+[diag(R _(i) ^(T))]⁻¹ N _(i)  Eq.(32)which implies crosstalk-free reception. The preferred MIMO precoderdescribed above corresponds to a single tone and is shown in FIG. 13.Combining the precoders of all tones and including the DMT transmittersand receivers, the vectored DMT system for downstream transmission isshown in FIG. 14. This system resembles the system of FIG. 12, exceptthat signals are “pre-processed” with precoders 1420-1 through 1420-Lbefore being sent by the system transmitters 1410-1 through 1410-L,respectively.

Assuming that transmit and receive filtering at the CO/ONU and at theCPE is identical, and noise within a tone has the same statistics forall users, the reciprocity property for twisted pair transmissionimplies that T_(i,up)=T_(i,down) ^(T). In that case, Equations (23) and(27) give the QR decomposition of the same matrix.

For the upstream channel, regardless of the loop topology, the diagonalelement of a column of T_(i) is larger in magnitude than theoff-diagonal elements of the same column. This occurs because, inupstream transmission, the crosstalk coupled signal originating from aspecific transmitter can never exceed the “directly” received signal ofthe same transmitter, and typically the magnitude difference is morethan 20 dB. The insertion loss of a signal is always smaller than thecoupling loss that it experiences when it propagates into a neighboringpair.

Visualizing the columns of T_(i) in vector space, it is seen that thecolumns are almost orthogonal to each other, which implies that Q_(i) isclose to being an identity matrix. Thus, the magnitudes of the diagonalelements of R_(i) do not differ significantly from those of the diagonalelements of T_(i), which indicates that QR cancellation performs almostas good as perfect crosstalk removal. This is illustrated in FIG. 15 fora two user case. As shown in FIG. 15, this holds for both possibledetection orderings.

The preceding discussion concerning upstream transmission can be readilyextended to downstream transmission by starting with the observationthat the crosstalk signals at a specific receiver can never exceed themagnitude of a “directly” received signal. Alternatively, the sameconclusions can be reached by using the transpose relationship betweenthe upstream and downstream channel matrices.

The computational cost incurred by the QR cancellation is decomposedinto the cost of the QR decompositions and the cost associated withsignal processing. DSL channels are stationary, so the QR decompositionsneed to be computed infrequently (preferably during initialization).Generally, the number of flops per tone (for example, using theHouseholder transform) can be greatly reduced by taking advantage of thecrosstalk environment characteristics. It is known that the crosstalknoise in a pair originates mostly from just three or four neighboringpairs, which implies that a typical T_(i) matrix is almost sparse withonly three or four relatively large off-diagonal elements per row.Therefore, approximating T_(i) as a sparse matrix, Givens rotations canbe employed to triangularize T_(i) with a reduced number of flops. Onthe other hand, the real-time computational burden due to the cancellerand precoder blocks cannot be reduced. In a straightforwardimplementation, the operations dominating the total cost are those ofEquations (24) and (28).

Although the assumption of perfect channel matrix knowledge isreasonable in the given environment, it is still worth brieflyconsidering the effects of channel estimation errors. The upstreamchannel matrix for a given tone can be estimated, including a channelestimation error. Then, the QR decomposition with the reciprocityassumption can be performed to get the QR factor estimates. Startingfrom Equation (24), the effect on upstream communication can becomputed. Doing this, it is found that the estimation errors impacttransmission by introducing a “bias” in the detection and also bypermitting some residual crosstalk. A similar analysis can be appliedfor downstream communication, but modulo arithmetic complicates theexpressions. Ignoring the modulo operations, the effects of theestimation errors can be separated into a detection bias term and aresidual crosstalk term.

The results of this analysis reveal that the impact of channelestimation errors is aggravated when any of the diagonal elements of{circumflex over (R)}_(i) are small. Although channel matrix singularityis almost impossible in the DSL environment, an ill-conditioned channel(implying small diagonal elements) cannot be ruled out, thus increasingthe impact of channel estimation errors and posing several computationalproblems. Such cases arise in high frequencies (for example, in looptopologies that have extreme loop length differences) or in the presenceof bridged taps. Nevertheless, the energy allocation algorithmsdiscussed below prevent the occurrence of such phenomena by not allowingtransmission in frequencies where the diagonal elements of R_(i) aresmall.

As seen above, the elimination of crosstalk in the signals of a systemwill substantially improve performance of the system. Optimizing energyallocations in the system, when taken in conjunction with the crosstalkelimination likewise improves system performance. Also, as noted above,appropriate energy allocation can help avoid problems resulting from theimpact of estimation errors in ill-conditioned channels.

Transmission Optimization

“Transmission optimization” as used in the following example will referto maximization of a weighted data rate sum. However, in the broadestsense of the present invention, the term “optimization” is not solimited. Optimization may also mean determining the maximum ratesavailable and allocating or provisioning available resources (includingdata rates for various users) within a digital communication system.

The methods disclosed in the following discussion concern energyallocation in frequency generally, energy allocation in frequency whileobserving constraints on induced crosstalk, and energy allocationcombined with upstream/downstream frequency selection.

Energy Allocation in General

The optimization objective is the maximization of the weighted sum ofthe data rates of all users:

$\begin{matrix}{\max{\sum\limits_{k = 1}^{L}{a_{k}R_{k}}}} & {{Eq}.\mspace{14mu}(33)}\end{matrix}$where α_(k)≧0 is the weight assigned to the k^(th) user, and R_(k) isthe achievable data rate of the k^(th) user, which may refer to eitherthe upstream or the downstream direction. In order to compute the datarate, an appropriate known gap approximation is employed. Taking intoaccount the fact that vectoring essentially “diagonalizes” the channel(and assuming no error propagation in the upstream direction), theupstream and downstream achievable rates are obtained:

$\begin{matrix}{R_{k,{up}} = {\sum\limits_{i \in N_{up}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{up}}^{i}}{\Gamma}} \right)}}}} & {{Eq}.\mspace{14mu}(34)} \\{R_{k,{down}} = {\sum\limits_{i \in N_{down}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{down}}^{i}}{\Gamma}} \right)}}}} & {{Eq}.\mspace{14mu}(35)}\end{matrix}$where Γ is defined as the transmission gap, and depends on theprobability of error requirement, the coding gain and the requiredmargin. Also, N_(up) and N_(down) are the sets of upstream anddownstream tone indices correspondingly, which depend on the FDD plan.Error propagation effects are generally limited since DSL systemsoperate at very small probabilities of error.

The parameters with respect to which optimization takes place areε_(k,up) ^(i) for upstream transmission and ε_(k,down) ^(i) fordownstream transmission. These parameters are constrained by limits onthe transmitted energy. In upstream transmission, the total transmitenergy is constrained by:

$\begin{matrix}{{\sum\limits_{i \in N_{up}}ɛ_{k}^{i}} \leq ɛ_{k,{up}}} & {{Eq}.\mspace{14mu}(36)}\end{matrix}$where ε_(k) ^(i) is the energy of (U_(i))_(k) in Equation (25), andε_(k,up) is the maximum allowed upstream transmitted energy of user k.Since ε_(k,up) ^(i)=ε_(k) ^(i), it is deduced that:

$\begin{matrix}{{\sum\limits_{i \in N_{up}}ɛ_{k,{up}}^{i}} \leq ɛ_{k,{up}}} & {{Eq}.\mspace{14mu}(37)}\end{matrix}$In downstream transmission, the total transmit energy constraint isexpressed as:

$\begin{matrix}{{\sum\limits_{i \in N_{down}}ɛ_{k}^{i}} \leq ɛ_{k,{down}}} & {{Eq}.\mspace{14mu}(38)}\end{matrix}$where ε_(k) ^(i) is the energy of (U_(i))_(k) in Equation (21), andε_(k,down) is the maximum allowed downstream transmitted energy of userk. Unfortunately, this constraint does not translate directly to aconstraint for {tilde over (ε)}_(k) ^(i)=ε_(k,down) ^(i), due tonon-linear precoding.

However, simulation results for extreme loop topologies indicate thatuse of the preferred precoder described above does not result inconsiderable correlation between the transmitted signals of differentusers. It is reasonable to assume that this result holds generally,since the simulated loops correspond to a worst-case situation withregard to the crosstalk coupling.

Therefore, the approximation ε_(k) ^(i)≈{tilde over (ε)}_(k,down) ^(i)is made and Equation (38) for downstream becomes:

$\begin{matrix}{{\sum\limits_{i \in N_{down}}ɛ_{k,{down}}^{i}} \leq ɛ_{k,{down}}} & {{Eq}.\mspace{14mu}(39)}\end{matrix}$

With this in mind, it is seen that the energy allocation problem ofEquation (33) becomes independent for each user, and thus the α_(k)weights are irrelevant in this scenario. The optimization problem foreach transmission direction is broken into k waterfilling problemsexpressed by:

$\begin{matrix}\begin{matrix}\max & {\sum\limits_{i \in N_{up}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{up}}^{i}}{\Gamma}} \right)}}} & {{Eq}.\mspace{14mu}(40)} \\{{subject}\mspace{14mu}{to}} & {{\sum\limits_{i \in N_{up}}ɛ_{k,{up}}^{i}} \leq ɛ_{k,{up}}} & {{Eq}.\mspace{14mu}(41)}\end{matrix} & \;\end{matrix}$and by:

$\begin{matrix}\begin{matrix}\max & {\sum\limits_{i \in N_{down}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{down}}^{i}}{\Gamma}} \right)}}} & {{Eq}.\mspace{14mu}(42)} \\{{subject}\mspace{14mu}{to}} & {{\sum\limits_{i \in N_{down}}ɛ_{k,{down}}^{i}} \leq ɛ_{k,{down}}} & {{Eq}.\mspace{14mu}(43)}\end{matrix} & \;\end{matrix}$Solutions to these problems can be derived using known techniques. Theresulting transmission spectra are optimal in the context of vectoredDMT.Energy Allocation with Power Back-Off

As discussed above, all users in the preferred vectored transmissioncorrespond to a group of neighboring twisted pairs. This does notpreclude the operation of other “alien” DSL systems in neighboringtwisted pairs, which on one hand cause crosstalk into the vectoredsystems, and on the other hand suffer from crosstalk originating fromthe vectored systems. The current approach in dealing with this problemis to impose limits on the transmitted power spectral densities (PSDs),so that the performance of systems is not excessively affected bycrosstalk.

Additionally, as in the situation illustrated in FIG. 5, VDSL systemssuffer from the fact that upstream signals on short lines detrimentallyaffect upstream performance on long lines (similarly to the near-farsituation in wireless communications). In order to avoid imposing anoverly restrictive universal PSD mask, power back-off methods have beenproposed which effectively make the PSD mask dependent solely on theloop length of the specific user. A similar scenario, where thedownstream communication of neighboring DSL systems may sufferconsiderably is shown in FIG. 16. Dramatic loop length differences willoccur more frequently as ONUs are installed on some lines while twistedpair connections to the COs remain.

Vectoring combined with full channel matrix knowledge can proveeffective in limiting the crosstalk induced by vectored systems, withoutresorting to the introduction of a universal PSD mask, or the use ofpower back-off methods (which do not necessarily take into accountknowledge about crosstalk coupling resulting from matrix channelidentification).

Equation (22) can be augmented to include the received samples of aliensystems:

$\begin{matrix}{\begin{bmatrix}Z \\Z_{n}\end{bmatrix} = {{\begin{bmatrix}T & C_{n} \\C & T_{n}\end{bmatrix}\begin{bmatrix}U \\U_{n}\end{bmatrix}} + \begin{bmatrix}N \\N_{n}\end{bmatrix}}} & {{Eq}.\mspace{14mu}(44)}\end{matrix}$where Z_(n), U_(n), and N_(n) are vectors of the received samples, ofthe transmitted symbols and of the noise samples, respectively, of thealien systems. The definitions of the block matrices C, C_(n) and T_(n)depend on both the channel and the characteristics of the alien DSLsystems; and, although T is block diagonal, this property will notgenerally hold for the other matrices.

When Z and Z_(n) correspond to systems belonging to different serviceproviders, it may be difficult to identify the crosstalk couplingmatrices C and C_(n), due to the fact that the current unbundlingframework does not allow the first provider to obtain access to either Zor Z_(n), and similarly for the second provider. However, a “thirdparty” of the type disclosed in U.S. Ser. No. 09/788,267, overcomes thisdifficulty by introducing an impartial third-party site or operation,which captures all transmitted and received data to produce estimates ofthe crosstalk coupling matrices.

Limiting the FEXT in the mean square sense, results in the followingconditions:

$\begin{matrix}{{{\sum\limits_{k = 1}^{L}{\sum\limits_{i \in N_{up}}^{\;}{{c_{j,{{{({k - 1})}N} + i}}}^{2}ɛ_{k,{up}}^{i}}}} \leq ɛ_{j,{up}}^{c}},\mspace{14mu}{j = 1},\ldots\mspace{11mu},{MN}_{N}} & {{Eq}.\mspace{14mu}(45)} \\{{{\sum\limits_{k = 1}^{L}{\sum\limits_{i \in N_{down}}^{\;}{{c_{j,{{{({k - 1})}N} + i}}}^{2}ɛ_{k,{down}}^{i}}}} \leq ɛ_{j,{down}}^{c}},\mspace{14mu}{j = 1},\ldots\mspace{11mu},{MN}_{N}} & {{Eq}.\mspace{14mu}(46)}\end{matrix}$where M is the number of neighboring systems, N_(N) is the number of“dimensions” (for example, the number of tones) per neighboring system,ε_(j,up) ^(c), ε_(j,down) ^(c) are the maximum allowable crosstalkenergies in sample j of the neighboring systems for upstream anddownstream, and c_(j,l) is the (j,l) element of the MN×LN matrix C. Notethat this approach can be generalized, so that both FEXT and NEXT arerestricted.

The set of inequalities in Equations (45) and (46), combinedcorrespondingly with those of Equations (37) and (39), form a set oflinear inequality constraints. Including the rate maximization objectiveof Equation (33) yields the following optimization problems:

$\begin{matrix}\begin{matrix}\max & {\sum\limits_{k = 1}^{L}{a_{k}{\sum\limits_{i \in N_{up}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{up}}^{i}}{\Gamma}} \right)}}}}} & {{Eq}.\mspace{14mu}(47)} \\{{subject}\mspace{14mu}{to}} & {{{\sum\limits_{i \in N_{up}}ɛ_{k,{up}}^{i}} \leq ɛ_{k,{up}}},\mspace{11mu}{k = 1},\ldots\mspace{11mu},L} & {{Eq}.\mspace{14mu}(48)} \\\; & {{{\sum\limits_{k = 1}^{L}{\sum\limits_{i \in N_{up}}^{\;}{{c_{j,{{{({k - 1})}N} + i}}}^{2}ɛ_{k,{up}}^{i}}}} \leq ɛ_{j,{up}}^{c}},} & {{Eq}.\mspace{14mu}(49)} \\\; & {{j = 1},\ldots\mspace{11mu},{MN}_{N}} & \;\end{matrix} & \;\end{matrix}$and by:

$\begin{matrix}\begin{matrix}\max & {\sum\limits_{k = 1}^{L}{a_{k}{\sum\limits_{i \in N_{down}}{\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{down}}^{i}}{\Gamma}} \right)}}}}} & {{Eq}.\mspace{14mu}(50)} \\{{subject}\mspace{14mu}{to}} & {{{\sum\limits_{i \in N_{down}}ɛ_{k,{down}}^{i}} \leq ɛ_{k,{down}}},\mspace{11mu}{k = 1},\ldots\mspace{11mu},L} & {{Eq}.\mspace{14mu}(51)} \\\; & {{{\sum\limits_{k = 1}^{L}{\sum\limits_{i \in N_{down}}^{\;}{{c_{j,{{{({k - 1})}N} + i}}}^{2}ɛ_{k,{down}}^{i}}}} \leq ɛ_{j,{down}}^{c}},} & {{Eq}.\mspace{14mu}(52)} \\\; & {{j = 1},\ldots\mspace{11mu},{MN}_{N}} & \;\end{matrix} & \;\end{matrix}$

The objective functions are concave (since they are sums of logfunctions), and the constraints form convex sets (because they arelinear inequalities). Thus, solutions can be efficiently produced usingknown convex programming techniques. Other restrictions (such as PSDmasks or bit caps) can be included in the above optimization problems,since they only require the introduction of linear inequalityconstraints, which preserve the convexity of the problem.

Energy Allocation and Upstream/Downstream Frequency Selection

Although all existing DSL systems employing FDD have a fixedupstream/downstream frequency duplexing band plan, a dynamicallyconfigured band plan may offer significant advantages. Such a plan iscommon for all users, but is determined during modem initialization,depending on the specific transmission environment, as well as userrequirements.

Examples of the disadvantages of a fixed band plan arise in the presenceof bridged taps, where transmission in one direction may face adisproportionate degradation, while transmission in the oppositedirection may remain unscathed. Adopting a dynamic band methodology insuch a case can provide a fairer distribution of the impact on bothupstream and downstream activity.

The optimization objective can now be expressed by:

$\begin{matrix}{\max\;{\sum\limits_{k = 1}^{L}\left( {{a_{k,{up}}R_{k,{up}}} + {a_{k,{down}}R_{k,{down}}}} \right)}} & {{Eq}.\mspace{14mu}(53)}\end{matrix}$where α_(k,up), α_(k,down)≧0 are the weights assigned to upstream anddownstream transmission for user k, and R_(k,up), R_(k,down) are theachievable upstream and downstream rates of user k. Here, theoptimization parameters involve not just the energies assigned but alsothe selection of upstream/downstream tones. However, if Equations (34)and (35) are used, the partition of the set of tones into N_(up) andN_(down) is a binary constrained problem, whose solution has very highcomplexity.

Instead, the binary constraints can be relaxed, which greatly simplifiesthe computations. This idea has previously been used for the computationof the FDMA capacity of the Gaussian multiple access channel in thepresence of intersymbol interference. In more detail, it initially isassumed that each tone is time-shared between upstream and downstream,thus obtaining the following achievable rates:

$\begin{matrix}{R_{k,{up}} = {\sum\limits_{i = 1}^{N}{t_{i,{up}}\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{up}}^{i}}{t_{i,{up}}\Gamma}} \right)}}}} & {{Eq}.\mspace{14mu}(54)} \\{R_{k,{down}} = {\sum\limits_{i = 1}^{N}{t_{i,{up}}\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{down}}^{i}}{t_{i,{down}}\Gamma}} \right)}}}} & {{Eq}.\mspace{14mu}(55)}\end{matrix}$where t_(i,up), t_(i,down) describe the fraction of time in tone i usedfor upstream and downstream transmission correspondingly, andt_(i,up)+t_(i,down)=1; t_(i,up)t_(i,down), t_(i,down)≧0. The existenceof t_(i,up) and t_(i,down) in the denominators inside the logexpressions implies that the assigned energy is “boosted” sincetransmission takes place over only some fraction of time. The energyconstraints for user k are:

$\begin{matrix}{{\sum\limits_{i = 1}^{N}ɛ_{k,{up}}^{i}} \leq ɛ_{k,{up}}} & {{Eq}.\mspace{14mu}(56)} \\{{\sum\limits_{i = 1}^{N}ɛ_{k,{down}}^{i}} \leq ɛ_{k,{down}}} & {{Eq}.\mspace{14mu}(57)}\end{matrix}$

Therefore, the optimization problem has the following form:

$\begin{matrix}\begin{matrix}\max & {\sum\limits_{k = 1}^{L}\left\lbrack {{a_{k,{up}}{\sum\limits_{i = 1}^{N}{t_{i,{up}}\frac{1}{2}\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{up}}^{i}}{t_{i,{up}}\Gamma}} \right)}}} +} \right.} & {{Eq}.\mspace{14mu}(58)} \\\; & \left. {a_{k,{down}}{\sum\limits_{i = 1}^{N}{t_{i,{down}}\frac{1}{2}{\log_{2}\left( {1 + \frac{{r_{k,k}^{i}}^{2}ɛ_{k,{down}}^{i}}{t_{i,{down}}\Gamma}} \right)}}}} \right\rbrack & \; \\{{subject}\mspace{14mu}{to}} & {{{\sum\limits_{i = 1}^{N}ɛ_{k,{up}}^{i}} \leq ɛ_{k,{up}}},\mspace{11mu}{k = 1},\ldots\mspace{11mu},L} & {{Eq}.\mspace{14mu}(59)} \\\; & {{{\sum\limits_{i = 1}^{N}ɛ_{k,{down}}^{i}} \leq ɛ_{k,{down}}},\mspace{11mu}{k = 1},\ldots\mspace{11mu},L} & {{Eq}.\mspace{14mu}(60)} \\\; & {{{t_{i,{up}} + t_{i,{down}}} = 1},\mspace{11mu}{i = 1},\ldots\mspace{11mu},N} & {{Eq}.\mspace{14mu}(61)}\end{matrix} & \;\end{matrix}$

The objective function is concave, because it is a sum of functions ofthe form

${x\;{\log\left( {1 + \frac{y}{x}} \right)}},$which are known to be concave in x,y≧0. The constraint sets are clearlyconvex, since they are defined by linear inequalities. Therefore, theproblem is convex and a variety of methods can be used to efficientlyderive a solution.

Still, such a solution would actually yield a hybrid between an FDD anda Time Division Duplexing (TDD) implementation. Since an FDDimplementation is required, an approximate solution is obtained byrounding t_(i,up) and t_(i,down). Naturally, this is suboptimal, butwhen the number of tones is fairly large, it will be adequately close tothe optimal solution. Simulation results validate this claim. Note thatthe power back-off constraints of the previous subsection also can beincluded in the problem formulation without considerably affecting thedifficulty of obtaining a solution.

In the above discussion, the objective has been the maximization of aweighted data rate sum. It will be apparent to one of ordinary skill,however, that by adjusting the weights, different surface points of thedata rate region achievable by vectored transmission can be determined,and thus the whole multi-dimensional surface can be determined as well.However, visualizing the inherent tradeoffs becomes difficult when theweighted sums include more than three terms. One practical question thatcan be posed to a service provider is whether a given vectored systemcan support a set of rate requirements and, if so, what energyallocation is required for achieving the requirements. This problemactually has a duality relationship with the weighted data rate sumproblem, and thus the weighted sum problem provides an alternativemethod to solve the “feasibility” problem.

Vectoring without power back-off or frequency planning can improveperformance significantly in several respects. For a given loop length,VDMT allows the achievement of much higher data rates. These rateincreases are considerable for lengths in the range of 3500-4500 feet orless. The gains can be even greater in short loops, where transmissionis obviously FEXT-limited. Also, vectored DMT can extend the maximumloop length given a data rate requirement. For example, a downstreamrate requirement of 50 Mbps typically limits a standard DMT system toloop lengths shorter than 1150 feet. Use of the present invention canextend the reach to lengths on the order of 2650 feet and possiblylonger.

The following table shows an example flowchart commensurate with theforegoing discussion. As illustrated in FIG. 17, one approach forcontrolling a digital communication system having a plurality ofdata-carrying communication lines having the available total power foruse in the system limited by a power constraint is implemented asfollows:

As depicted by block 1700, the total power constraint for each line isassigned an initial value. As depicted by block 1702, a competitivelyoptimal data rate is determined for each line. According to one exampleembodiment, the competitively optimal data rate is determined byperforming the steps of: determining a power allocation within the totalpower constraint of each line by iteratively allowing each line tooptimize its power allocation as detected in block 1701, and determiningthe competitively optimal data rate for each line based on thedetermined power allocation for the line in block 1702. As depicted byblock 1703, a model of the line, signal and the actual interferencecharacteristics of the communication lines is created. As depicted byblock 1704, signals are processed using the model to remove interferencefrom signals including evaluating the competitively optimal data ratefor each line. According to one example embodiment, the signals areprocessed by performing the steps of: comparing the competitivelyoptimal data rate of a line with a target rate for the line as depictedin block 1705; increasing the power constraint for a line if thecompetitively optimal data rate of the line is less than the target ratefor the line as depicted in block 1706; decreasing the power constraintfor the line if the competitively optimal data rate of the line exceedsthe target rate for the line by at least a prescribed variance asdepicted in block 1708; maintaining the power constraint for the line ifthe competitively optimal data rate of the line is equal to the targetrate for the line; and maintaining the power constraint for the line ifthe competitively optimal data rate of the line exceeds the target ratefor the line by less than the prescribed variance as depicted in block1707.

When energy allocation is further constrained by the requirement thatsome “alien” DSL system also must be protected against crosstalk fromthe vectored system, then a service provider can perform power back-offin a “selective” way, so that the performance impact can be distributedaccording to the service priorities. Further improvements are seen whenthe upstream/downstream duplexing frequency plan is allowed to vary ineach vectored bundle depending on the loop characteristics and theservice requirements.

Generally, embodiments of the present invention employ various processesinvolving data transferred through one or more computer systems and/ormodems. Embodiments of the present invention also relate to a hardwaredevice or other apparatus for performing these operations. Thisapparatus may be specially constructed for the required purposes, or itmay be a general-purpose computer selectively activated or reconfiguredby a computer program and/or data structure stored in the computer. Theprocesses presented herein are not inherently related to any particularcomputer or other apparatus. In particular, various general-purposemachines may be used with programs written in accordance with theteachings herein, or it may be more convenient to construct a morespecialized apparatus to perform the required method steps. A particularstructure for a variety of these machines will be apparent to those ofordinary skill in the art based on the present description.

In addition, embodiments of the present invention relate to computerreadable media or computer program products that include programinstructions and/or data (including data structures) for performingvarious computer-implemented operations. Examples of computer-readablemedia include, but are not limited to, magnetic media such as harddisks, floppy disks, and magnetic tape; optical media such as CD-ROMdisks; magneto-optical media; semiconductor memory devices, and hardwaredevices that are specially configured to store and perform programinstructions, such as read-only memory devices (ROM) and random accessmemory (RAM). The data and program instructions of this invention mayalso be embodied on a carrier wave or other transport medium. Examplesof program instructions include both machine code, such as produced by acompiler, and files containing higher level code that may be executed bythe computer using an interpreter.

The many features and advantages of the present invention are apparentfrom the written description, and thus, the appended claims are intendedto cover all such features and advantages of the invention. Further,since numerous modifications and changes will readily occur to thoseskilled in the art, the present invention is not limited to the exactconstruction and operation as illustrated and described. Hence, allsuitable modifications and equivalents are deemed to fall within thescope of the invention.

1. An arrangement for controlling communication in a digitalcommunication system having a plurality of communication lines on whichsignals are transmitted and received, the signals being affected byinterference during transmission, the arrangement comprising: a firstprocessing arrangement that creates a model of the interferencecharacteristics of a communication line selected from the plurality ofcommunication lines, the interference characteristics due to signalscarried on the plurality of communication lines; a second processingarrangement that processes signals using the model to determineinterference characteristics created by actual signals carried on theother communication lines and in response, remove interference from theactual signals; and a plurality of user blocks, wherein at least oneblock operates independent from other blocks and adapted to collectinformation about line, signal and interference characteristics of thecommunication lines at a common processing node responsive to the actualsignals, wherein each user block is permitted to transmit and receivesignals using a data rate and wherein the second processing arrangementis further adapted to analyze a weighted sum of the data rates of thesignals over the plurality of communication lines.
 2. The arrangement ofclaim 1, wherein the second processing arrangement further allocatesenergy to each of the plurality of communication lines for transmissionof signals.
 3. The arrangement of claim 2 wherein the signals are sentusing a plurality of frequencies and further wherein the secondprocessing arrangement is further adapted to dynamically adjust thefrequencies used to send the signals.